There are many applications that use fast switching, overshoot free current sources, especially though not exclusively in communications and digital data transmission systems, full motion color display video applications, opto-isolators drivers, infrared light emitting diode (LED) communication devices operating at high data rate, general purpose LED drivers in devices with or without serial interface, and in display devices where the light intensity is current dependent. In view a prominent importance among the numerous applications of fast switching, overshoot free current sources, the ensuing description may exemplarily refer to the driving of an electrical load in the form of an LED, though other equivalent electrical loads may be similarly driven.
FIG. 1 shows a basic LED driver circuit suitable for monolithic multi-channel drivers for LED panel displays, a partial block diagram of which is shown in FIG. 2. Light output is a function of current IOUT; by changing IBIAS, which has a ratio K with IOUT, it is possible to modulate the intensity level. The “reference” and the “sensing” (feedback) resistors may be of the same-type and well-matched.
In an integrated circuit (IC), the biasing current IBIAS is usually the result of a processing/amplification (e.g.: 1:1) of an input current, generated by the user on an external resistor, coupled to a suitable pad and biased by a temperature and supply compensated voltage reference (typically a Band Gap reference). The output current is thus temperature and supply independent and a DMOS, if technologically available, is often employed as a power output element.
The MOS MGSW (FIG. 1) acts as a switch and grounds the gate of the power element, thereby preventing it from remaining floating when the driver is disabled (ENABLE=0). In these conditions the op-amp has its input and output terminals to zero voltage.
When the driver is enabled (ENABLE=1), supposing the positive input rises instantaneously, the op-amp has to raise its output (i.e. the gate of the power) from zero to at least the threshold voltage of the DMOS (in a worst scenario, up several hundreds mV, when operating at the internal supply voltage value). The op-amp negative input may be increased (usually from few tenths of mV to several hundreds mV) to the appropriate value: VSENS=VREF, for setting the output current to the design value.
Passing from one situation to another, not in “small signal” conditions, the dynamic response of the system is basically conditioned by the slew-rate of the op-amp. Slew rate (SR) is related to the dominant pole of the open loop amplifier and to the charging current of the gate capacitance (including the Miller capacitance). The most general kind of operational amplifier is depicted in FIG. 3 and by definition:
      SR    =                            [                                    ⅆ              Vo                                      ⅆ              t                                ]                max            =                        I                      O            ⁢                                                  ⁢            1                                    C          C                      ,  andCC is the capacitance needed to introduce a dominant pole to compensate the op-amp. Remembering that
                    f        T            =                        gm          ⁢                                          ⁢          1                          2          ⁢                      πC            C                                ,                  ⁢    and              SR      =                                    I                          O              ⁢                                                          ⁢              1                                            gm            ⁢                                                  ⁢            1                          ⁢        2        ⁢        π        ⁢                                  ⁢                  f          T                      ;  slew rate can be increased by increasing the transition frequency fT value and/or the saturation current IO1 of the first stage or by decreasing the gm1 of the same stage.
Many drivers may be able to switch high currents (for example, 80 mA, 100 mA, 500 mA) and this usually calls for the use of large output transistors (Power-DMOS) that have large feedback parasitic capacitance (CGD), which in turn appears multiplied by the gain of the output stage (gm*RL) of the driver and increases with diminishing drain voltage, affecting the dynamic performances of the circuit.
In LED panel displays applications, the LED brightness is usually controlled by adjusting the output constant current, set by mean of an external resistor; moreover “dimming” is often used and comprises switching ON/OFF the current at high rate (a switching frequency of few MHz may be used). If a 5 MHz dimming is implemented (with a 50% duty-cycle), the driver is used to have a rise time much shorter then the 100 ns half period.
An output setup time, for example, less then 20 ns, may be needed at least to improve the performance of the system. If the simple architecture of FIG. 1 is used, very high performance in terms of GBW and slew rate would be demanded of the Op-Amp in order to meet with the specifications.
High slew-rate and bandwidth provides for high bias currents, a relatively complex design for the Op-Amp, high large power consumption and high silicon area consumption, especially in multi-channel devices (to be noted that 16 channels are very frequently used). It is also known to resort to additional support circuitry to improve the speed of the driver.
As known, “one-shot” circuit may be used, as depicted in FIG. 4, for providing a suitable amount of current in a pulsed way; this may help in charging the gate of the power DMOS in a very short time. There remain several potential drawbacks and limitations in these known techniques for fast switching current driving of loads such as a LED: 1) The switching performance of known circuits are strongly dependent on: the output current level; the electrical characteristics of the load LED (i.e. its equivalent RC circuit); the size of the output power element (dictated by current capability specifications); and the bandwidth and slew rate characteristics of the Op-Amp. 2) Under the same output current (IOUT) conditions, if the circuit may drive LEDs of many different characteristics, a large spectrum of resistive loads may be considered in the equivalent circuit: by dimensioning the system to match the rise time specifications for the higher values of load resistance (worst case), it may exhibit unacceptable current spikes at lower load resistance values; and because of Miller's multiplication effect, the gate capacitance increases with the load resistance, moreover the CGD increases with the consequent lower drain voltage.
3) Under identical resistive load conditions, speed performance is greatly dependent on the output current level to be set. Because of the different levels of gate voltages that are requested at different currents, there may be a risk of not matching all the specifications because if the device may provide for a wide range of currents to be set, it is not simple to match the speed requirement at, for example, 80 mA and the current spikes constraint at 3 mA (as a matter of fact, the one-shot current could be “too low” in the first case and “too high” in the second one). The circuit would need additional circuitry to modulate and control the “energy” of the “one-shot” circuit on the basis of the set level of the output current.
4) Under same resistive load and output current conditions, the rise time is dependent on the external supply voltage VLED. In fact, as it is well known, the parasitic capacitance CGD is inversely proportional to the VDS voltage value. For this reason, even if the charge current (energy) is modulated in dependence of the output current, the overshoot in the output current increases with VLED.
The problem with the “one-shot” technique may be the difficulty to control the gate charging process in all load and IOUT−VLED conditions. Often the gate voltage and hence the output current exhibit high spikes that can reach 50% or even more of the final value of the set output current. On the other hand, expedients to reduce the spike (the quantity of current charging the gate and/or the duration of the pulse) may slow-down the device, risking not meeting the speed requirements. A difficult trade off is generally sought between speed and current spike issues.
In U.S. Patent Application Publication No. 2008/0012507 to Nalbant, a variety of techniques for fast switching through high brightness and high current LEDs using current shunting devices are disclosed. The disclosed techniques may be burdensome to implement in multi-channel devices, e.g. 16 channels, because of large silicon area and power consumption in view of the fact that the shunting device may be sized to divert the full load current.
U.S. Pat. No. 6,346,711 to Bray describes a technique to improve the response time that makes use of additional current feed components to the LED during its illumination phase. The additional large size switches and related control circuitry (all switches may carry the maximum design current) increase, significantly the silicon area and power consumption.
U.S. Pat. No. 6,144,222 to Ho discloses a high speed programmable current driver used for infrared LED communication devices. Large area critical precision requirements in a multi channel device may be burdensome. U.S. Pat. No. 6,469,405 to Moya et al. discloses a technique to reduce overshoot issues. Also this technique uses additional switches in the output current path, which may be suitably sized for the maximum design current at minimum voltage drop condition.
U.S. Pat. No. 6,734,875 to Tokimoto et al. and U.S. Pat. No. 6,288,696 to Holloman are other publications dealing with LED displays. In the latter, a technique is disclosed to control the current driving by an analog voltage set by an analog drive line including a sample and hold circuit. Drivers designed, for example, for full color full motion video applications, often use internal pulse width modulation (PWM) controls, which give the capability to increase the visual refresh rate and to reduce flickering effects, thereby improving fidelity.
This, together with the need to suitably modulate the brightness of the LEDs, could make the driver output capable of being switched ON/OFF at high rates (according to this technique, the “ON” period can be scrambled into several short “ON” periods). Indeed, pulse widths as short as 30 ns could be requested and the driver circuit may be fast enough to set the current at a stable level within such pulses of extremely short width.
In any case, it is always of paramount importance to reduce as much as possible and ideally prevent any switching spike produced by fast switching circuits such as drive current source circuits. This avoids damage to a driven load as a LED, power dissipation (specially in case of a multi-channel device simultaneously switching array LEDs) and EMI issues. Moreover, for securely dealing with very short pulses, it is important to control intensity and duration of the spike, in order to avoid appreciably varying the mean value of the current (e.g. the brightness within the illumination phase of a driven LED).